Frequency VCO Design and Schematics
Iulian Rosu, YO3DAC
This note will review the process
by which VCO (Voltage Controlled Oscillator) designers choose their oscillator’s
topology and devices based on performance requirements, components types and DC
Basic oscillator design
specifications often require a given output power into a specified load at the
design frequency. The drive level and bias current set the fundamental output
current and the oscillation frequency is set by the resonator components.
Transistor selection of the
transistor should consider noise, frequency, and power requirements. Based on
the particular device, the design may account for parasitics of the device
affecting resonator components as well as nonlinear performance specifications.
All the VCO schematics presented
below were practical build using the Infineon SiGe transistor BFP420, and any of
them can be re-tuned for different frequency ranges changing varicaps and LC
The VCO must exhibit a low Phase Noise in
order to meet the Sensitivity, Adjacent Channel and Blocking requirements. In
digital modulation scheme the VCO’s Phase Noise affects the Bit Error Rate
requirements. High Pushing (change of the oscillation frequency with
supply voltage) can cause Phase Noise degradation due to increased sensitivity
to the power supply noise.
Phase Noise varies typically by
3dB with temperature, in the –55ºC to +85ºC range.
A buffer at the output is necessarily to
isolate the VCO from any output load variations (Pulling) and to
provide the required output power. Meeting simultaneously the output power and
load pull specification directly with a stand-alone oscillator would be
difficult. However, this buffer amplifier requires a higher supply current.
Alternative would include to use at the output circulators, isolators or
VCO output power is usually measured into a 50
ohm load. Output power requirements specified in dBm, and tolerances vs tuning
frequency in ± dB.
The tuning slope is the slope of the frequency
to voltage tuning characteristic at any point and is the same as modulation
sensitivity. The slope could be positive or negative. For a positive slope,
the output frequency. increases as the tuning voltage increases. Similarly for
a negative slope, the output frequency decreases as the tuning voltage
A monotonic tuning characteristic means that
the frequency is single valued at any tuning voltage and that the slope has
the same sign across the tuning range.
as a function of tuning voltage is a measure of tuning linearity. For
any given application, have to specify the minimum and maximum of the tuning
sensitivity. In the case of a VCO, the frequency coverage is rather restricted
since the influence of the feedback network is small compared to the active
device itself. Conventional oscillator designs (with a LC circuit or
transmission-line equivalent coupled to a negative-resistance active device
will only provide a restricted frequency coverage and poor stability). A
negative resistance can easily be obtained from most microwave transistors
when considering chip and package parasitics.
Tuning flatness -
As the VCO frequency
range is increased, the difficulty to achieve a flat output power is
increased. Adding an output filter to suppress harmonics may in some
power output flatness.
The drive level should consider the trade-off between
harmonic content, oscillator stability, and noise.
In order to lower
the VCO Phase Noise, a number of rules should be respected:
The active device has noise properties which
generally dominate the noise characteristic limits of an oscillator. Since all
noise sources, except thermal noise, are generally proportional to average
current flow through the active device, it is logical that reducing the
current flow through the device will lead to lower noise levels.
Narrowing the current pulse width in the active
device will decrease the time that noise is present in the circuit and
therefore, decrease Phase Noise even further.
Maximize the loaded Q of the
tuned circuit in the oscillator.
There is a trade-off between the
Q factor of the oscillator, its size and its price. The low Q-Factor of an LC
tank and its component tolerances needs careful design for phase noise without
individual readjustment of the oscillators.
Usually a larger resonator will
have a higher Q (e.g. a quarter wavelength coaxial resonator).
Choose an active device that
has a low flicker corner frequency.
A bipolar transistor biased at a
low collector current will keep the flicker corner frequency to a minimum,
typically around 6 to 15 KHz (Most semi-conductor manufacturers can provide the
frequency corner (fc) of their devices as well as the 1/f characteristic.
Maximize the power at the
output of the oscillator.
In order to increase the power at
the input of the oscillator, the current has to be increased. However, a low
current consumption is critical to preserving battery life and keeping a low fc.
In a practical application, the current will be set based on output power
required to drive the system (typically a mixer), and then the Phase Noise will
need to be achieved through other means.
Choose a varactor diode with a
low equivalent noise resistance.
The varactor diode manufacturers do not measure or
specify this parameter. The best approach is then empirical; by obtaining
varactors from several vendors and experimentally finding out which one yields
the lowest phase noise in the VCO circuit and thus has the lowest equivalent
There are two basic types of varactors: Abrupt and
- The abrupt tuning diodes will
provide a very high Q and will also operate over a very wide tuning voltage
range (0 to 60 V). The abrupt tuning diode provides the best phase noise
performance because of its high quality factor.
- The hyperabrupt tuning diodes,
because of their linear voltage vs. capacitance characteristic, will provide a
much more linear tuning characteristic than the abrupt diodes. These are the
best choice for wide band tuning VCO's. An octave tuning range can be covered in
less than 20 V tuning range. Their disadvantage is that they have a much lower Q
and therefore provide a phase noise characteristic higher than that provided by
the abrupt diodes.
Keep the voltage tuning gain (Ko)
to the minimum value required.
This is the most challenging
compromise because the thermal noise from the equivalent noise resistance of the
varactor works together with the tuning gain of the VCO to generate phase noise.
This compromise will be the limiting factor determining the phase noise
Noisy power supplies may cause additional noise.
Power supply induced noise may be seen at offsets from 20 Hz to 1 MHZ from the
carrier. If the VCO is powered from a regulated power supply, the regulator
noise will increase depending upon the external load current drawn from the
regulator. The phase noise performance of the VCO may degrade depending upon
the type of regulator used, and also upon the load current drawn from the
regulator. To improve the phase noise performance of the VCO under external
load conditions it is always a good design philosophy to provide RF bypassing
of power and DC control lines to the VCO. RF chokes and good bypassing
capacitors (low ESR) is recommended at the DC supply lines. This will minimize
the possibility of feedback between stages in a complex subsystem. Improved
bypassing may be provided by incorporating an active filter
Parallel Tuned Colpitts VCO
There are 3 types of BJT Colpitts
VCOs. Common-Collector, Common-Emitter and Common-Base.
The most used is Common-Collector
configuration where the output is often taken from the collector terminal,
simply acting as a buffer for the oscillator connection at the base-emitter
This is the only Colpitts
arrangement in which the load is not part of the three-terminal model or the
oscillator equation; though care must be taken to ensure that the collector
output voltage does not significantly feedback through the base-collector
As an alternative, the output of
the common collector could also be taken across emitter resistance Re.
The ratio of the feedback capacitors in the Colpitts
VCO (C3 and C4), is more important than the capacitor’s actual values. A good
place to start is with a one to one ratio. The loaded Q of the resonator
circuit can be increased by reducing C3 or increasing C4. Doing so however,
reduces the loop gain in the oscillator, and enough loop gain must be
maintained to guarantee oscillation start-up under all conditions (mainly
under different temperatures and system output loads).
The value of the collector resistor, R3 affects the
oscillator loop gain. As in a common collector amplifier, the lower the
impedance in the collector circuit the more loop gain the circuit will have.
This resistor provides another means of controlling the loop gain of the
oscillator since a good oscillator design has just enough loop gain to
guarantee reliable oscillation start-up. If there is to much loop gain the
oscillator will operate in deep compression which will load the Q of the
resonator circuit because the input impedance at the base of the transistor is
very low when current saturation occurs. The resistor also tends to minimize
the level of the harmonics.
L2 is chosen as an RF choke to provide a high
impedance in the emitter circuit and ensure that most of the oscillator power
is fed back to the base of Q1 instead of being dissipated in R2.
Emitter resistor R2 is used for current feedback thus
providing a stable DC bias point that will be independent of the beta of the
C1 capacitor defines the amount of coupling between
the active device and the resonator. The lighter the coupling (a smaller value
of C1), the better the loaded Q of the resonator is, which results in a better
phases noise performance. However, the compromise is a reduced output power
and the potential for the VCO not to start under all operating conditions
(especially at higher temperatures when current gain is reduced). Designing
the system with too light of a coupling may also results in a sensitive design
which may yield potential manufacturing problems.
The final tuning component of the oscillator, C2 sets
the voltage tuning gain of the oscillator. This capacitor should keep the
coupling as light as possible while maintaining the required frequency tuning
range of the VCO so that the varactor’s phase noise contribution is reduced to
a minimum. If the coupling is too light, the oscillator may not start under
certain conditions. The worst case condition for this oscillator topology is
when V-varicap is set at zero volts.
A good way to check if C2 is large enough for
reliable oscillator start up is to monitor the output power of the VCO with
zero volts on the tune line. The power with V-varicap at 0V should be within 1
dB of the power with V-varicap at 3V. If C2 is too small, the output power of
the VCO will fall off sharply when V-varicap approaches zero volts or the
oscillator may stop completely.
One good reason to use a transistor with a high Ft
such as the BFP420 (Ft = 25GHz) is that C2 can be small and oscillation
start-up will be reliable simultaneously.
In order to ensure that the loaded Q of the resonator
circuit is not the limiting factor in phase noise performance, the varactor
can be replaced with a fixed 2.5pF capacitor and compare the results. A
varactor can degrade up to 5-6dB
The varactor can reduce the Q of the resonator
circuit but this effect is secondary to the varactor modulation due to its own
equivalent noise resistance. One way of reducing this effect is to parallel
two or more varactors of smaller value while keeping the same tuning curve.
This effectively reduces the equivalent noise resistance.
Series Tuned Colpitts VCO (Clapp
The series-tuned Colpitts circuit
(or Clapp oscillator) works in much the same way as the parallel one.
The difference is that the variable capacitor, C1, is
positioned so that it is well-protected from being swamped by the large values
of C3 and C4.
In fact, small values of C3, C4 would act to limit
the tuning range. Fixed capacitance, C2, is often added across the varicap to
allow the tuning range to be reduced to that required, without interfering
with C3 and C4, which set the amplifier coupling.
The series-tuned Colpitts has a reputation for better
stability than the parallel-tuned original. Note how C3 and C4 swamp the
capacitances of the amplifier in both versions.
oscillation frequency is given by: ω2 L =
Wideband Colpitts VCO
This wideband Colpitts VCO uses a series back-to-back
connection of two SMV1232 varactors instead of a single varactor. This
connection allows lower capacitance at high voltages, while maintaining the
tuning ratio of a single varactor. The back-to-back varactor connection also
helps reduce distortion and the effect of fringing and mounting capacitances.
The wideband Colpitts feedback capacitances C3, C4
were optimized to provide a flat power response over the wide tuning range.
These values may also be re-optimized for phase noise if required.
The circuit is very sensitive to the transistor
choice (tuning range and stability) due to the wide bandwidth requirement.
DC bias is provided through resistors R6 and R7,
which may affect phase noise, but allows the exclusion of RF chokes. This
reduces costs and the possibility of parasitic resonances which is the common
cause of spurious responses and frequency instability.
The Hartley VCO is similar to the parallel tuned
Colpitts, but the amplifier source is tapped up on the tank inductance instead
of the tank capacitance. A typical tap placement is 10 to 20% of the total
turns up from the “cold” end of the inductor. (It’s usual to refer to the
lowest-signal voltage end of an inductor as cold and the other, with
the highest signal voltage as hot.). The same as in Colpitts case a
good place to start is with a one to one ratio.
C2 limits the tuning range as required.
C1 is reduced to the minimum value that allows
reliable starting. This is necessary because the Hartley’s lack of the
Colpitts’s capacitive divider would otherwise couple the transistor
capacitances to the tank more strongly than in the Colpitts, potentially
affecting the circuit’s frequency stability.
Wideband Differential Push-Push
The circuit schematic shows a pair of transistors in
a single feedback loop, connected so that collector currents would be 180°
shifted. A pair of back-to-back connected SMV1232 varactors is used to allows
lower capacitance at the high voltage range, without changing the tuning
biasing is provided through resistors R8, R9 and R10, which may affect the
phase noise, but eliminate the need for inductive chokes. This eliminates the
possibility of parasitic resonances that could affect the wide tuning range
and also cause for frequency instability and spurs.
The DC chokes, L1 and L2 are used for phase
correction between pairs and their losses is dominated by the series emitter
resistors R6 and R7.
The DC blocking series capacitances C1 and C2,
including associated parasitics, shall have the SRF outside of the tuning
A three-pole Low Pass Filter at the output is used to
filter the unwanted spurious.
Differential Cross-Coupled VCO
The cross-coupled differential
transistor pair presents a negative resistance to the resonator due to positive
This negative resistance cancel
the losses from the resonator enabling sustained oscillation.
Frequency variation is achieved
with two varicap diodes BB135.
An optimal trade-off between
thermal noise- induced phase noise and DC power dissipation can be achieved
when the oscillation amplitude is designed to set the differential pair
transistors to operate at the boundary between saturation and linear regions.
The excess noise factor F is dominated by the noise
from the tail current source near even harmonics of the carrier frequency. In
order to improve phase noise this contribution has to be minimized. An
efficient way of doing this is to use a noise filtering technique. An inductor
L3 and capacitor C5 forms a 2nd order low-pass filter which prevents noise at
even harmonics from being injected into the feedback path of the oscillator.
The noise filter leaves low-frequency noise from the
tail current source unaffected. Low-frequency noise from the tail current
source is also up-converted to the carrier as amplitude modulation.
Low-frequency noise on the tuning line modulates the non-linear capacitance of
the varactors giving rise to phase noise variation with control voltage.
The phase noise degradation due to control voltage
noise is very significant at the lower tuning range where the varactors are
most non-linear. The stack of two varactors reduces the varactor gain Kvco
at the lower tuning range which in turn reduces phase noise variation with
Negative Resistance VCO
The resonator of the Negative Resistance VCO is a
series-tuned base network consisting of two series varicap capacitances and an
inductor for the positive reactance element.
Performance is highly dependent on the transistor
type. Certain component values are critical.
This oscillator actually works best when lower Ft
transistors are used. The circuit can be envisioned as a series-tuned Clapp,
with internal transistor base-to-emitter capacitance and collector-to-emitter
capacitance acting as a voltage divider. Microwave transistors with little
internal capacitance do not work well except at the high end of the useful
range of this oscillator type. Higher Ft devices required increased
capacitance added at the emitter. At the low end of the frequency range,
adding external base-to-emitter capacitance is sometimes necessary,
If bias conditions result in a emitter resistance
below about 200 ohms, an RF choke may be required in series with the
resistance. This choke must be free of any resonances in the operating
The output can be taken from several points. The L1
inductor can be tapped. As the tap is moved toward the transistor, more power
is coupled out. If the tap is too close to the transistor, the loading reduces
the oscillation margin, and the operating frequency becomes more load
The output can be taken by
capacitive coupling at the emitter (low level) or at the collector (higher
level, but have more spurious).
Because the negative resistance oscillator uses a
series-tuned resonator, the varactors lead inductance becomes a part of the
resonator. This is an advantage over varactor-tuned oscillators using parallel
resonators. The base coupling capacitor inductance and transistor base
inductance are also absorbed.
The loaded Q of negative resistance oscillators is
typically less than 5 and this circuit defies attempts at improving the Q.
When used as a broadband varactor-tuned VCO, the low loaded Q does not limit
phase noise performance significantly because varactor modulation noise
predominates, particularly at higher offset frequencies.
Franklin oscillator uses two
transistor stages having the same common terminal (emitter for bipolar device)
when the greater power gain and better isolation from the resonant circuit is
possible compared with the case of a single-stage configuration.
There are two possible
configurations for the resonant circuit, parallel and series. The circuit presented below uses
a parallel LC resonant circuit (L1 and the varctor diode).
In the case of a parallel
resonant circuit configuration, the resonant LC circuit is isolated from
the input of the first stage and the output of the second stage by means of
small shunt capacitances C1 and C2 having high reactances at the resonant
In this circuit, each stage
shifts phase 180° so
that the total phase shift is 360°
which is equivalent to zero phase shift. We may say that one stage serves as the
phase inverting element in place of the RC or LC network which generally
performs this function. It is, from an analytical viewpoint, immaterial which
stage we choose to look upon as amplifier or phase inverter. The configuration
is essentially symmetrical in this respect; both stages provide amplification
and phase inversion.
The salient feature of the Franklin oscillator is
that the tremendous amplification enables operation with very small coupling to
the resonant circuit.
Therefore, the frequency is relatively little
influenced by changes in the active device, and the Q of the resonant circuit is
substantially free from degradation.
The closest approach to the high frequency stability
inherent in this oscillator is attained by restriction of operation to, or near
to, the Class-A region. This should not be accomplished by lowering the
amplification of the two stages, but, rather by making the capacitors C1 and C2
Additionally, a voltage-follower 'buffer' stage is
helpful in this regard. Extraction of energy directly from the resonant tank,
would, of course, definitely negate the frequency stability otherwise
Obviously, the Franklin oscillator is intended as a
low-power frequency-governing stage, not as a power oscillator.
The Goral VCO has an
emitter-follower stage inserted in the feedback path of an otherwise
conventional Colpitts oscillator circuit.
The midpoint of the capacitive divider (which is
actually part of the resonant tank) now sees a much lower impedance with respect
to ground than would be the case without the emitter follower.
Because the feedback gain of a Goral VCO is greater
than a standard Colpitts, placing a feedback resistor R*, helps improving the
overall phase noise performance and improving also the linearity, reducing the
VCO generated harmonics.
The power gain of the JFET/BJT combination is much
greater than that of the JFET 'oscillator' alone. There is latitude for
considerable experimentation in the ratio of the two capacitors used in the
Colpitts section of the circuit. This ratio can be optimized for frequency
stability without easily running out of feedback.
Note that the emitter-follower is
directly coupled to the JFET. It may be necessary to experiment with
bias-determining resistances to ascertain Class-A operation from the
emitter-follower. Also, the output transistor is intended to operate in its
To provide higher isolation of
the load from the VCO resonant circuit a cascode VCO configuration, can be used.
The negative resistance
oscillation conditions for common emitter transistor Q1 are provided by using
the feedback inductance L1.
And here is the winner. If you
want to build a very stable, low phase noise, and low spurious VCO, definitely
Vackar VCO is the choice.
This is not a common type in the
RF “professional” world, one reason could be the name of its inventor.
A Vackar VCO is a variation of
the split-capacitance oscillator model. It is similar to a
Colpitts or Clapp
VCO in this respect. It differs in that the output level is more stable over
frequency, and has a wider
compared to a Colpitts or Clapp design.
The Vackar VCO circuit
incorporates a π-section
tank to attain the needed 180°
phase-reversal in the feedback loop.
However, the inverted feedback signal is not
directly fed back to the input of the active device; rather, it is loosely
coupled through a small capacitor. Often, a shunt capacitor is introduced to
further reduce the coupling.
The basic idea is to isolate the resonant circuit as
much as possible from the input of the active device, consistent with obtaining
This circuit is particularly advantageous with
solid-state devices, and especially with bipolar transistors that have
inordinately-low input impedances and that present a widely-varying reactance to
the tuned circuit as a consequence of temperature and voltage changes.
Once the overall circuit is operational, the values
of capacitance C1 in series with Cvar and collector capacitance
(C2) may be optimized for best stability. Generally, it will be found that the
capacitor closest to the collector of the transistor can be several times larger
than the capacitor associated with the base circuit.
The introduction of attenuation in the feedback loop
(via the small capacitor in the Vackar) prevents over-excitation and effectively
isolates the resonant circuit from the active device.
The frequency tuning range of Vackar VCO is above one octave, not
observable to many oscillator types.
The frequency tuning is provided independently of the
coupling to the LC tank circuit.
The parametric variables of the transistor (which
depends by the bias current and temperature), are isolated from the resonator.
The transistor input is not overloaded as other VCO
circuits and the collector output has low impedance providing low gain just to
maintain the oscillation.
The feedback division ratio is fixed (typical range
for coupling ratio is 1:4 up to 1:9). Even if the VCO is tuned, the impedance
divider is fixed, in this way increasing the stability.
Two negative sides of Vackar VCO are the critical
starting oscillation point, and the low output level, which always requires to
use a buffer amplifier.
oscillation doesn’t start means that it doesn’t have enough positive feedback,
as for to begin the oscillation and maintain it in the time. In the above
schematic C3 and C4 are critical values finding this point.
used as an RF choke with SRF outside of the frequency range, to don’t affect
the tuning range and flatness over frequency. It
is important that the RF choke in the collector circuit 'looks good' at the
operating frequency (presents a high inductive reactance). Resonances from
distributed capacitance in the choke windings, especially those in the
series-resonant mode, can degrade stability or even inhibit oscillation.
Ferrite-core chokes are generally suitable for this application. Sensitivity to
RF choke characteristics is common to all oscillator circuits that use chokes
for shunt-feeding the DC operating voltage to the oscillator.
1. Alpha Industries - VCO
2. Minicircuits - VCO Application
3. Oscillator Basics and
Low-Noise Techniques for Microwave Oscillators and VCOs - U.Rohde
4. Oscillator Design and Computer
Simulation - R.Rhea
5. RF and Microwave Transistor
Oscillator Design – A. Grebennikov
6. Practical Oscillator Handbook - I. Gottlieb
7. RF Design Magazine - 1997 -
Microwave Journal - 1997 - 2008
9. Microwaves & RF - 2002 - 2006