Best regards,
Dr. Howard Johnson
At 10:02 PM 6/11/98 -0400, you wrote:
>Dr. Johnson,
>
>Thanks for your reply. I agree with you that measuring (and simulating) the
>power ground impedances on a complex board is not an easy task. In a
>real-life situation it is complicated by the fact that boards are not
>designed with probe points for this purpose, therefore attaching the probes
>may be extremely difficult. I also agree with your derivation which
>illustrates how much noise you would pick up with adjacent loops in a
>typical situation. This clearly shows that we need to go an extra distance
>to achieve our goal. Your derivation also suggests the direction where the
>solution can be found: the only help is to reduce the mutual inductance,
>since reducing the source voltage would reduce dI/dt but would also reduce
>the voltage to be measured by the same ratio. By using smaller loops
>(closer connection points between the shield and signal wire, and very short
>exposed signal and shield connections), the 'noise floor' of the measurement
>can be reduced to about 25 milliohms up to about one GHz, which may be
>enough today for some of the boards. This noise floor can be achieved even
>if the connection points are just 0.1 inches away, which brings up another
>interesting point: on a power-ground distribution system we want to know not
>only how much noise will propagate from one device to the other, but also it
>is important to figure out how much noise a particular device is generating
>at its own location. Having the full impedance matrix (or any other
>electrical matrix for that matter) gives the possibility to sum up the noise
>contributions from several devices at the desired locations. To have the
>self-impedance measured (Z11), we clearly need to connect the source and the
>probe so close that at our measurement frequencies the electrical separation
>is negligible. By using semirigid for both the source cable and probe cable
>and using very short interconnections, we were able to measure self and
>transfer impedances and to correlate to simulated values within less than
>one dB error up to 100MHz and within less than 3dB error up to one GHz. To
>achieve the good correlation it is also necessary to use the proper complex
>impedances in the equivalent schematic of voltage divider.
>
>But finally let me repeat again that I agree with you, it is not an easy
>task.
>
>Best regards
>
>Istvan Novak, SI Engineer
>SUN Microsystems
>-----Original Message-----
>From: Howard Johnson <howiej@sigcon.com>
>To: Istvan NOVAK <istvan.novak@worldnet.att.net>
>Cc: si-list@silab.Eng.Sun.COM <si-list@silab.Eng.Sun.COM>
>Date: Tuesday, June 09, 1998 6:22 PM
>Subject: Re: [SI-LIST] : Re: MEASURING POWER GROUND IMPEDANCE
>
>
>>NOTE: Since this question was copied to the S.I. list reflector,
>>I will copy my answer there as well. I of course always welcome
>>any other comments on the subject, pro or con. Measuring the
>>power and ground impedance on a complex board is never easy.
>>
>>
>>Hello Istvan,
>>
>>Regarding the comment below about "polluting your measurement",
>>the issue is that we are trying to measure a very small signal (the
>>Vcc noise voltage), and any direct crosstalk between the signal
>>source and the probe may overwhelm the signal you are trying to
>>measure.
>>
>>Example: suppose we have a 50-ohm sinewave source, and a 50-ohm probe.
>>Adjust the sinewave source so we see 1V p-p when the source is plugged
>>straight into the probe (no power system connected).
>>The total driving point impedance at the probe point is 25 ohms (fifty
>>from the source in parallel with fifty from the probe).
>>Now connect both probe and scope to the power system on your bare
>>board (no active parts installed - just the bypass caps).
>>If the Vcc-ground impedance is 0.1 ohms (a reasonable value for a good
>>power system), then the Vcc-ground impedance will reduce the voltage
>>received at the scope by a ratio of (0.1)/(25) -- that's the
>>resistor divider theorem. (In the analysis below I just converted this
>>number to a dB figure.)
>>The voltage at the scope will now be about 4 mV. This is a small,
>>but clearly visible and easily measured amount of noise.
>>
>>Now let's look at the direct coupling between the source and the probe.
>>Couple both source and probe to the board using coax (I
>>like to use RG-174, because it's thin, flexible, and easy to solder
>>to the board). Let's say that at the source attachment point, the
>>distance between the coax shield attach point (to board ground) and
>>the coax signal attach point (to board Vcc) is about 1/2 inch,
>>and that the coax signal conductor stands up about
>>1/8 inch above the surface of the board. The total exposed area of this
>>loop between the signal conductor and the ground conductor
>>is (1/2)*(1/8) = 0.0625 square inches.
>>Assume the same for the probe connection.
>>
>>If the source and probe coax cables are connected to points
>>separated by about 1 inch, the mutual inductive coupling between
>>these two loops will be on the order of 0.02 nH. The total
>>di/dt flowing through the source loop, times the mutual inductance
>>between loops, will coupled crosstalk noise voltages directly
>>into the probe loop.
>>
>>At a frequency of, say, 500 MHz, the total noise voltage
>>will be: Lmutual*(di/dt = 0.02nH*2*(pi)*500MHz*(1V/25ohms) = 2.5 mV
>>
>>The noise voltage is almost as large as the signal you are trying to
>>measure. Keep the source and probe cables searated by at least
>>an inches, and keep the length of the exposed coax signal conductor
>>as short as practical.
>>
>>Also, at frequencies in the multiple-megahertz range and beyond,
>>you will notice that the impedance is a function of the
>>position of the source and probe cables, due to various resonance
>>effects between the power and ground planes. If you're interested
>>in that topic, check out the cool simulations from SIGRITY (and probably
>>many others) on the subject.
>>
>>
>>Best regards,
>>Dr. Howard Johnson
>>
>>
>>
>>
>>At 08:08 AM 6/1/98 -0400, you wrote:
>>>Dr. Johnson,
>>>
>>>I am following with great interest the On-line Newsletters of High-Speed
>>>Digital Design. Inj one of your recent responses regarding the
>measurement
>>>of power/ground planes, you are suggesting not to connect the probes to
>>>nearby points.
>>>Can you elaborate somewhat further your reasoning for that ('local
>crosstalk
>>>will pollute your measurement')?
>>>Also, can you give some more reasoning behind your suggestion of using the
>>>resistance divider theorem in this case? Do you mean here the magnitudes
>of
>>>impedances?
>>>
>>>Thank you
>>>
>>>Istvan Novak, SI Engineer, SUN Microsystems
>>>
>>>-----Original Message-----
>>>From: High-Speed Digital Design Mailing List <hsdd@accessone.com>
>>>To: High-Speed Digital Design Newsletter Subscriber(4) : ; <High-Speed
>>>Digital Design Newsletter Subscriber(4) : ;>
>>>Date: Tuesday, May 26, 1998 4:28 PM
>>>Subject: MEASURING POWER GROUND IMPEDANCE
>>>
>>>
>>>>*---------------------------------------------------------------*
>>>> H i g h - S p e e d D i g i t a l D e s i g n
>>>>
>>>> *On-Line Newsletter*
>>>>
>>>> Dr. Howard Johnson, Vol. 2 Issue 14
>>>>*---------------------------------------------------------------*
>>>>
>>>>*------------------------(ANNOUNCEMENTS)------------------------*
>>>>
>>>>Next Public High-Speed Digital Design Seminars:
>>>>
>>>> U. of Oxford, UK June 22-23, 1998
>>>> San Jose, CA September 21-22, 1998
>>>>
>>>>Registration & information at:
>>>>http://signalintegrity.com/seminar.htm
>>>>
>>>>Tell Your Co-Workers!
>>>>
>>>>*--------------------------(QUESTION)---------------------------*
>>>>
>>>>MEASURING POWER GROUND IMPEDANCE
>>>>Larry Smith of Sun Microsystems writes:
>>>>
>>>> Dr. Johnson - thanks for your newsletter, I have just
>>>> subscribed. I completely agree with the comments pertaining
>>>> to power plane impedance and the 'single node' assumption
>>>> below the frequency where the board resonates (Volume 2 Issue
>>>> 14). We have checked the impedance of power planes with a
>>>> network analyzer. With no capacitors present, you can see
>>>> interesting resonances that depend on the 1/4 wavelength from
>>>> the probes to the card edge (valleys) and half wavelengths
>>>> that fit into the card dimensions. But these measurements
>>>> always come in dB and my spice simulations come out in ohms
>>>> (after I force 1 Amp). Do you have any ideas on how to
>>>> correlate between them? I would like to be able to measure
>>>> the plane impedance in Ohms!
>>>>
>>>> A minor point... We are looking at using HIGHer dielectric
>>>> constants for the material between power planes in order to
>>>> gain more decoupling capacitance. That is going to lower the
>>>> resonant frequencies of the power planes, possibly into
>>>> frequencies of interest (near the clock).
>>>>
>>>> Best Regards,
>>>>
>>>>*-------------------(REPLY FROM DR. JOHNSON)--------------------*
>>>>
>>>> Thanks for your interest in High-Speed Digital Design.
>>>>
>>>> When using a network analyzer to measure power and ground
>>>> impedance, we drive the power and ground planes at one point
>>>> on the board with a sine wave, and the measure how much
>>>> voltage appears at a different point.
>>>>
>>>> Don't set the IN and OUT cable attachment points on the board
>>>> too close together or else their local crosstalk will pollute
>>>> your measurement.
>>>>
>>>> In this setup, we can relate ohms to dB if (1) we know the
>>>> driving point impedance of the network analyzer test setup
>>>> and (2) the network analyzer is calibrated in terms of dB
>>>> gain, where 0 dB means there is no device under test
>>>> connected (that is, the IN cable is directly connected to the
>>>> OUT cable).
>>>>
>>>> To establish this relation, we just use the resistance
>>>> divider theorem:
>>>>
>>>> dB gain = 20*log(Vmeasured / Vreference)
>>>>
>>>> dB gain = 20*log( Rpwr-gnd / (Rpwr-gnd + Rsource) )
>>>>
>>>> where Rsource is the driving point impedance of the test
>>>> setup.
>>>>
>>>> If your network analyzer has a 50-ohm output, and a 50-ohm
>>>> input, then the driving point impedance at the device-under-
>>>> test point is 25 ohms. Rsource = 25 ohms.
>>>>
>>>> Now we can simplify things a little if we use the fact that,
>>>> even when resonating, the power and ground planes have an
>>>> impedance much less than 25 ohms (if they didn't the whole
>>>> system wouldn't even come close to working). Therefore we can
>>>> ignore the term Rpwr-gnd in the denominator, and just use the
>>>> approximation:
>>>>
>>>> dB gain = 20*log( Rpwr-gnd / Rsource )
>>>>
>>>> Converting this formula to express Rpwr-gnd in terms of dB,
>>>> we get:
>>>>
>>>> Rpwr-gnd = Rsource *{10 <raised to the power of> [(dB
>>>> gain)/20]}
>>>>
>>>>
>>>>Best regards,
>>>>Dr. Howard Johnson
>>>>
>>>>*---------------------------------------------------------------*
>>>>
>>>> Comments welcome! hsdd@signalintegrity.com
>>>>
>>>> Newsletter Archives:
>>>> http://signalintegrity.com/newsletter.htm
>>>>
>>>
>>_________________________________________________
>>Dr. Howard Johnson, Signal Consulting, Inc.
>>tel 425.556.0800 fax 425.881.6149 email howiej@sigcon.com
>>
>>High-Speed Digital Design seminars:
>>June 22-23 at Oxford U., in the U.K.,
>>September 21-22 in San Jose, CA
>>S E E - - - >>> WWW.sigcon.com
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>
>
_________________________________________________
Dr. Howard Johnson, Signal Consulting, Inc.
tel 425.556.0800 // fax 425.881.6149 // email howiej@sigcon.com
http://WWW.sigcon.com -- High-Speed Digital Design books, tools, and workshops
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