It's nice to have the ability to reduce the output power from an
amplifier. One way to do this is to reduce the anode-voltage and
anode-current simultaneously so that the output load-R of the
amplifier tube or tubes does not change appreciably. This allows the
tank circuit to function at its design Q for both high and low
power.
Switching primary taps is not an efficient method of reducing output
voltage because in order to do so extra turns must added to the
primary. To make room for the extra turns, the primary's wire
diameter must be decreased--and that increases R. An efficient method
of reducing the DC output voltage in a HV power supply is by
switching secondary taps on the transformer. If a fairly ordinary
ceramic rotary switch is insulated from the chassis, it can easily
perform this job. The taps should not be switched under load.
If no secondary tap is provided on a transformer, it is possible to
lower the output voltage 50% by switching from fullwave doubler to
fullwave bridge rectification. All that's needed is a suitable SPST
vacuum relay, or well-insulated ceramic switch, two filter capacitors
and four strings of rectifiers. For example, a power supply that
produces 4000V for SSB could be operated at 2000V for RTTY, CW, or
FM. The DC output current capability doubles when the output voltage
is halved during fullwave bridge operation--just what's needed for
FM's and RTTY's much higher duty-cycle. When switching the voltage
output, it is best to temporarily switch the power supply off and
then restart. The output voltage may be switched down without
switching the supply off--provided that the amplifier is in standby.
On paper, the variable auto-transformer, a.k.a. Variac® or
Powerstat®, looks good. Variacs/Powerstats are intended to be
used with resistive loads. When a Variac is set at or near 100% of
the input voltage, it adds only a small amount of series R. However,
when a Variac is set to produce a fraction of the input voltage it
adds more series R. This is of little consequence with resistive
loads. However, when the load is a capacitor filter DC supply, due
the demand for high peak current, additional series R is most
unwelcome. Although Variacs perform acceptably with resonant choke
filter power supplies, using a Variac to control the output voltage
from a capacitor filter supply is not good engineering practice.
A Variac can be used in place of a step-start relay. Provided that
the operator always remembers to set the Variac to near-zero before
switching the amplifier on, all will be well. A step-start relay
offers some advantages: it is cheaper, mistake-proof, saves many
kilograms, and it adds substantially less series R.
There is, however, an appropriate step-start application for a
Variac. Eimac® recommends using a motor-driven Variac, feeding
the filament transformer primary, to bring up and bring down the
filament-voltage (over a period of two minutes each) on its tubes
which incorporate water-cooled filament supports. An example is the
8973 tetrode--just what you need if you are building a 600kW linear
amplifier.
Most transformers use paper to separate and insulate each layer of windings. Paper is hygroscopic--i.e., it absorbs water vapour from the air. The presence of water reduces the insulating ability of the paper. In time, insulation breakdown is likely. The solution is to pot the windings. Plastic resins are best. Petroleum tar is next best. Since potting fills up the air spaces in the windings--and air is a poor heat conductor--potting also improves heat transfer--thereby reducing internal temperature and increasing MTBF. Potting adds very little to the initial cost of a transformer and subtracts substantially from the long-term cost. Some custom transformer manufacturers offer potting as an extra-cost option. Peter Dahl Co. has a potting option.
Commercial transformer potting is normally done in a vacuum
chamber to facilitate the evacuation of air bubbles. However, with a
little patience, it is possible to pot transformers satisfactorily
without special equipment. Bake the transformer in an oven at a
temperature of about 175 degrees F/80 degrees C. Bake for two to
three hours per pound. Baking drives out internal moisture. After
baking, place the transformer on a table covered with a thick layer
of newspapers. Position the transformer so that the leads or lugs are
down. Using masking tape, seal the end of the transformer windings
opposite the leads/lugs so that liquid can not escape easily when the
transformer is inverted.
Polyester fiberglass laminating resin is designed to flow into small
spaces and expel air bubbles. It can be used for potting
transformers.
In a clean tin-can, pour in a quantity of laminating resin that will
fill up the air spaces in the bottom 5% of the transformer's
windings. Using pliers, bend the rim to facilitate pouring the resin.
Mix in about 5 drops of catalyst per ounce of resin. Depending on the
ambient temperature and humidity, this amount of catalyst will result
in a moderately fast gel time. Pour the resin slowly into the
windings. Resin pouring should be done steadily and from only one
area of the windings to avoid trapping air bubbles. Any leaks from
the bottom can be patched by forcing raw silicone rubber into the
area of the leak. When the resin gels, it forms a thin bottom plug.
The bottom plug need not be more than about 5mm thick.
Pour an amount of resin into the can that will fill the remainder of
the windings. For a several-kVA transformer, use about 1.5 drops of
catalyst per ounce of resin. For smaller transformers, slightly more
catalyst is needed. The resin must not gel before the air bubbles
have had a chance to escape--so it is better to err on the light side
for the amount of catalyst.
Heat increases the fluidity of the resin--hastening the exit of
bubbles. However, heat tends to decrease gel-time. Internal
transformer heating is accomplished by forcing current through the
windings with a Variac. Connect the Variac to the highest voltage
winding. Short the highest current winding with an AC ampmeter.
Increase the voltage until the ampmeter indicates the rated winding
current. At this level fairly normal internal heating results. As
soon as the resin begins to gel, stop the current and direct a
cooling fan at the transformer. Resin-gelling is an exothermic
reaction.
The most frequent failure mechanism for HV power supply rectifiers
is too much reverse current. This problem can be virtually eliminated
in 50Hz/60Hz, fullwave bridge and fullwave doubler, capacitor filter
circuits if the total PIV in each string of diodes exceeds the
no-load DC output voltage by at least 50%. For operation in
high-temperature environments, a 100% factor may be needed.
Modern solid-state rectifiers are made differently than they were 30
years ago. In those ancient times, same-type rectifiers did not have
uniform reverse characteristics. Rectifier failure was common. In an
attempt to compensate for the inherent weaknesses in early
solid-state devices, rectifier protection schemes were used.
Resistors and capacitors were paralleled with series
rectifiers--probably a take-off on the practice of using equalizer
resistors on electrolytic filter capacitors. However, in any series
circuit, the currents in all of the elements are exactly equal. Thus,
when rectifiers are in series, the reverse current burden is exactly
the same for each rectifier--provided that no parallel resistors are
used. Manufacturers of series rectifier units long ago abandoned the
practice of using parallel resistors and capacitors. The 1995/6/7
Radio Amateur's Handbook explains why rectifier 'equalization' is
prone to cause premature rectifier failure.[page 11-9, middle column,
top]
Series-connected rectifiers should be of the same type. Mixing
rectifiers types in the same series string could cause a problem
during the reverse half-cycle.
When a rectifier has been conducting, it takes a finite amount of
recovery time for the rectifier to stop conducting after the source
of forward current reaches zero. It is important that a rectifier not
be conducting when the reverse voltage arrives. This can be a problem
when rectifying high frequency AC or when rectifying square waves.
Paralleling a capacitor with each rectifier may help the rectifiers
to stop conducting sooner. If you need to rectify high frequency AC,
one solution is to use fast-recovery epitaxial rectifiers. 1000PIV,
1A, 70 nanosecond recovery time units are currently priced at about
50¢ each in quantities of 100.
Rectifiers that have a rating above 1kV PIV are typically made from a series of individual rectifiers that are entombed in an epoxy package. This arrangement makes for a neat-appearing installation--but there is a trade-off. Epoxy is a poor conductor of heat. Individual series-connected diodes mounted on perfboard and exposed to open air dispose of heat much more efficiently than do multi-diode packages.
Filter capacitors usually have a ripple-current rating. The
ripple-current rating should be at least equal to the maximum DC
output current capability of the supply. Quality filter capacitors
are designed to minimize equivalent series resistance [ESR]. Low ESR
ohms translates into a high ripple-current rating.
Oil-filled capacitors are available in two types: filter service, for
use in power supplies, and flash service for use in photography or
pulsed laser applications. The flash capacitor is designed for
maximum capacitance per unit volume. To reduce volume, very thin
metal foil is used to make the plates of a flash capacitor. Thin
plates have more ESR--so they dissipate more I-squared times R power
when they are subjected to ripple-current.
For longest life in high duty-cycle applications, cool air should be
allowed to circulate freely around filter capacitors.
According to some manufacturers, flash capacitors can be used in
filter service if they are operated at 60% of their rated peak volts.
In intermittent duty applications, it may be possible to use flash
capacitors at more than 60% of their rated peak-voltage. To discover
how a flash-capacitor is faring in ripple-current service, after
about an hour of contesting, if the capacitor is warm to the touch,
an internal heating problem is indicated. Internal heating causes
expansion and stresses the capacitor's case--which may eventually
come apart at the seams and begin to leak dielectric oil.
If not plainly stated on its label, there is a way of determining the
intended type of service for an oil-filled capacitor.
Flash-capacitors usually have a peak voltage [PV or VP] rating.
Filter-service capacitors are usually rated in DC working volts
[DCWV]. Capacitors can also be rated in AC working volts. To convert
AC-working volts to DC-working volts, multiply the AC voltage by
three.
There have been instances where surplus flash capacitors were offered
for sale with altered or counterfeit markings. For example, a
capacitor that was originally marked "3.5kVP" was changed to "5kV" by
erasing characters. Thus, a capacitor that should have been de-rated
to 0.6 x 3.5kV = 2100V for filter service would appear to be good for
5kV. A practical way of determining whether an oil-filled capacitor
can withstand ripple-current is to connect it in series with an
AC-ampere meter and an AC voltage source. The voltage is adjusted
until the AC current is equal to the expected maximum output current
of the power supply. If, after an hour, the capacitor shows little or
no internal heating, you have a winner.
There is also a flash service type of aluminum electrolytic
capacitor, that is not designed to handle ripple-current.
Electrolytic filter capacitors are intolerant of reverse current and
heat. Electrolytic capacitor working voltage [WV] ratings should be
treated with respect. The WV rating is virtually the maximum voltage
rating. Despite their more delicate nature, electrolytic filter
capacitors offer substantial advantages over oil-filled filter
capacitors. The main advantages are more joules of energy storage per
dollar, reduced weight and reduced volume.
When electrolytic capacitors are operated in series, they should
share the voltage equally. In order to do this, a voltage equalizer
resistor is connected across each capacitor. Equalizing resistors
must have fairly equal resistance--and their resistance should not
change appreciably during aging. If an equalizer resistor changes
value appreciably, domino-effect destruction of an entire section of
filter capacitors may result.
There is no formula for determining the optimum resistance for an
equalizer resistor. Less resistance equals less bleed-down time.
However, less resistance produces more heat. A compromise is in
order.
Carbon-composition resistors change resistance with age. This
characteristic is unacceptable for equalizer resistor service. High
resistance, wire wound resistors are wound with extremely fine
resistance wire. They are not remarkably reliable. Metal oxide film
[MOF] resistors are more reliable. The initial resistance of a MOF
resistor is typically much closer to the labeled value--and it will
stay that way for many years. A Matsushita/Panasonic® 3W, 100k
ohm MOF resistor makes a good equalizer resistor for 450V capacitors.
It produces a reasonable bleed-down time and a reasonable amount of
heat. These resistors are available from Digi-Key.
Electrolytic filter capacitors are ruined quickly by reverse current.
Reverse current often occurs when a rectifier fails. To protect
electrolytic capacitors from reverse current, connect a >600PIV
diode across each capacitor. The cathode band of the diode connects
to the capacitor's positive terminal.
When a grounded-grid amplifier's operating bias is obtained from a
single Zener diode, there is no way to compensate for tube variation.
One solution is to obtain the operating bias from a series string of
forward biased rectifier diodes. By switching the number of diodes in
and out with a rotary switch, the bias can be changed in
approximately 0.7V increments.
Traditionally, a mechanical relay has been used to switch amplifier
bias between receive and transmit. An optoisolator coupled to a
transistor switch, i.e., an electronic bias switch, can do this job
faster, more reliably, sans-noise, and cheaper.
There are principally two means of actuating electronic bias
switches--RF-actuation and coil-current actuation. Although it sounds
hip, RF-actuation creates two problems. The amplifier rapidly
switches between linear bias and non-linear bias during softly-spoken
syllables of speech. This causes choppy-sounding audio and splatter.
When the electronic bias switch is controlled by the current that
passes through the RF relays' coils, it is not possible to
intermittently switch the amplifier into non-linear bias during
transmit. Coil current actuated bias switching can be accomplished
with an optoisolator. The optoisolator's input LED is driven by the
coil-current. The output of the optoisolator drives the bias switch
transistor.
Since the grid draws virtually zero current, it is easy to make the bias continuously adjustable in Class AB1 operation. Typically, the cutoff bias voltage during receive will be about 50% higher than the transmit bias voltage. An optoisolator driving a HV FET can be used to switch the bias between transmit and receive. A circuit is provided.
A conventional relay switches in roughly 25mS. Such relays have traditionally been used for RF and bias switching in RF amplifiers. This was acceptable when transceivers also used conventional relays. Currently manufactured transceivers are designed for AMTOR, QSK telegraphy, and unobtrusive SSB VOX operation. Modern transceivers T/R and R/T switch quietly, and do so in as little as 5mS. Such radios typically use a high-power SPDT reed relay to switch the antenna between transmit and receive. Similar relays can be used for amplifier input RF switching. Jennings and Kilovac manufacture high speed, SPDT vacuum-relays that have a continuous rating of 7A at 32MHz [2450W into 50 ohms]. The Jennings relay is the RJ-1A. Kilovac's relay is the HC-1. When used with a speedup-circuit, either relay can switch in under 2mS. Although both manufacturers make DPDT RF vacuum-relays, none are as speedy as their fastest single pole models. Thus, separate input and output relays are usually faster than a single DPDT relay.
Another device that can be used for high-speed RF switching is the
PIN [P-Intrinsic-N] diode. PIN diodes are similar to 1000PIV
rectifier diodes--i.e., they have a wide intrinsic region. PIN diodes
are utilized extensively in radars as transmit-receive switches.
A PIN diode is switched off by applying DC reverse voltage to widen
its intrinsic region. The PIN diode is switched on by passing DC
current in the forward direction to fill its intrinsic region with
current carriers. PIN diodes are extremely fast switches. Their
lifetime is virtually unlimited as long as the allowable PIV is not
exceeded.
The typical reverse breakdown voltage rating for a PIN diode is
around 1000V. A legal-limit amateur radio amplifier produces an
output voltage of about 800 volts peak-to-peak [p-p] into a 50 ohm
load--so a 1000PIV PIN diode is more than adequate. When the load Z
is higher, due to a somewhat less than wonderful SWR, the switching
device may be exposed to more than 1000Vp-p. This poses no problem
for a typical high-speed vacuum-relay. Even if a vacuum-relay's
breakdown voltage is temporarily exceeded, there is little likelihood
that permanent damage to the vacuum-relay's contacts will result.
However, solid-state devices are not so forgiving. A single
voltage-transient can destroy a PIN diode.
For 100 WPM computer-CW, the PIN diode is clearly the only choice.
For 30 WPM CW, AMTOR and high-speed VOX, a vacuum-relay has
advantages.
Different types of solid-state components are rated somewhat
differently. Some ratings are realistic. Some ratings are not
realistic. The maximum ratings of large transistors and large Zener
diodes can not be realized unless drastic, extreme measures are used
to keep the case temperature below the maximum allowable 25 degrees C
at full ratings. In the real world, operation at 30% of a published
dissipation or current rating is usually safe. Additionally, bipolar
power transistors suffer from a generic weakness called
secondary-breakdown phenomenon. For example, a "1500V, 8A, 150W"
power transistor may only be able to safely dissipate 15W at moderate
collector-to-emitter voltages. T-MOS power FETs are much more
resistant to secondary-breakdown.
Wire-lead rectifier current ratings are fairly realistic when they
are mounted on perfboard and cooling air is allowed to circulate
freely around individual rectifiers.
There is some variation in the inverse breakdown voltage of solid-state rectifiers of the same type. Measuring the breakdown voltage of each rectifier diode is a good precaution. When a diode's reverse current reaches 1 to 2 uA, the voltage across the diode is the breakdown voltage. Exceeding this voltage is likely to be fatal. As operating temperature increases, breakdown voltage decreases.
A breakdown voltage tester (a.k.a. high-pot) could be described as
a variable HV Ohmmeter that does not read directly in ohms. It is a
useful tool. Breakdown testers are essential for testing vacuum
relays, vacuum capacitors, blocking capacitors, air-variable
capacitors, rectifiers, and for finding problems with insulation.
Building or troubleshooting a tube-type RF amplifier without a
breakdown tester is like crossing an ocean without a navigation
instrument. For most amateur radio applications, the highest voltage
component rating commonly encountered [with vacuum-capacitors and
vacuum-relays] is 15kVDC/9kV RF peak, so a 0 to 15kV breakdown tester
should suffice.
Although commercial breakdown testers are available, they are not
inexpensive. A suitable breakdown tester can easily be constructed
from mostly-surplus parts. The main parts are a 50/60Hz low-current
HV transformer, a >1A, 120V variable transformer, a 120V
incandescent bulb, some diodes, resistors, a sensitive uA meter and
two HV filter capacitors.
Commercial, low-current HV DC supplies may also be used provided that
they are connected to a Variac in series with a 120V incandescent
light bulb to limit current. The bulb limits the short-circuit
current to a safe value--obviating the need for a fuse. The wattage
of the bulb is roughly proportional to the wattage of the supply. The
rated [I=P/E] bulb current should be similar to the appropriate fuse
rating for the HV supply primary. A multi megohm resistor is used to
limit the current flow into the device under test. The uA meter
should be protected with back to back 1A diodes. A circuit is
provided.
Most power supplies benefit from something to soften the shock of
start-up. A 10A DPST-NO or 10A DPDT relay and two approx..25 ohm 10W
resistors are just about all that's needed to add a step-start
circuit to the average 1500W amplifier. The step-start circuit goes
in series with the mains fuses or circuit breakers. With this
arrangement the filaments, the HV supply and the LV supplies enjoy
the benefit of a kinder and gentler start-up.
Every HF amplifier has at least two resonant circuits in its
output circuitry. The more obvious one is the HF-resonant tank
circuit. A less obvious one is the VHF-resonant circuit that is
principally formed by the anode capacitance and the inductance of the
conductors between the tank circuit and the anode. In 1500W
amplifiers, anode resonance typically occurs around 100MHz--well
within manufacturers' ratings for "Amplifier and Oscillator Service"
for the tubes that are commonly used in such amplifiers.
The equivalent resistance of a high Q parallel resonant circuit is
virtually infinite. A low Q parallel resonant circuit has a
relatively low equivalent resistance.
The voltage gain of an amplifier tube is roughly proportional to the
load resistance. High load resistance produces more gain. Low load
resistance produces less gain.
If the conductors in the anode resonant circuit have a high VHF Q,
the equivalent load resistance presented to the anode will be high
and the tube will exhibit increased voltage gain at the VHF
resonance. If the conductors in the anode resonant circuit have a low
VHF Q, the load resistance presented to the anode will be low and the
voltage gain at the VHF anode resonant frequency will be reduced. Of
course, if no VHF energy were present, it would make no difference
how much VHF gain an HF amplifier had.
When a transient current passes through a resonant circuit, the
resonant circuit rings like a bell--producing a damped sine wave
signal. This is how ancient spark transmitters produce RF--and the
larger ones produced many kilowatts of it.
Whenever the anode-current in an HF amplifier changes, a small VHF
damped sine wave signal is produced in the anode's VHF resonant
circuit. This signal can be observed with a VHF oscilloscope or a
spectrum analyzer. The amplitude of the RF voltage produced is
proportional to the Q of the anode resonant circuit. If none of this
damped wave signal were fed back to the input, there would be no
problem.
In a grounded-grid amplifier, the grid appears to shield the input
from the output. In a grid-driven Class AB1 amplifier, The
RF-grounded screen appears to shield the input from the output.
However, no grid and no screen is perfect--so some of the damped wave
VHF signal at the anode is capacitively fed back to the input--and
amplified.
Although it's unlikely, if the phase and amplitude of the damped wave
signal happens to be just right, oscillation at the anode's VHF
resonance can occur. If the VHF energy that is produced could find
its way to a load, no danger would be posed by a VHF oscillation.
However, the HF tank circuit is a low pass filter that effectively
blocks VHF energy. Thus, the oscillator is unloaded and the resulting
grid-current is very high. The unloaded condition can cause VHF
voltage transients in the anode circuit. These transients may cause
tune-capacitor arcing and band switch arcing across open contacts.
Since they are closest to the anode resonant circuit, open
tune-capacitor padder contacts, as well as open 10m contacts are most
vulnerable to parasitic-instigated arcing. Band switch contacts can
be melted and/or vapourized by such occurrences.
On page 72, the 1926 Edition of The Radio Amateur's
Handbook tells us how to build an improved VHF parasitic
suppressor The logic was elementary. A suppressor is supposed to
dampen the anode circuit. Since low Q is synonymous with high
dampening, why not decrease Q by using resistance-wire? Quoting from
page 72:........ "The combination of both resistance and
inductance is very effective in limiting parasitic oscillations to a
negligible value of current."
After 1929, someone forgot to include this information in the
Handbook. In those days, the oversight probably didn't matter very
much. Large amplifier tubes generally had low VHF amplification, so
VHF instability was not a major issue. During the ensuing decades,
people got into the habit of using parasitic suppressors made from
copper or silver-plated copper. This was an easy habit to get into
since copper and silver can be soldered more easily and cheaply than
nickel/chromium [nichrome] resistance wire. Meanwhile, the
performance of amplifier tubes kept improving. Because of these
improvements, modern high-amplification tubes appear to benefit more
from 1926-vintage low VHF-Q parasitic suppressors than 1926-vintage
tubes. NOTE: In 1926, a 'high-mu' triode had a mu of around 40.
Low VHF-Q conductor material can be used to increase VHF loading
in the anode resonant circuit. Nickel-chromium-iron alloys are best,
Nickel-chromium (nichrome) and some types of stainless-steel are
almost as good. The use of copper, aluminum, and silver should be
kept to a minimum. However, good conductors are desirable beyond the
tune capacitor, which marks the end of the anode VHF resonant circuit
and the beginning of the HF tank circuit.
Output Z: The output impedance of most tubes is a matter of
kilo-ohms--not ohms. There is no scientific reason to use "heavy
duty" conductors between the anode and the tune capacitor. If good
VHF stability is a design goal, it's best to use conductors that are
no larger than is necessary to carry the highest current present,
i.e., the 10m RF circulating current between the anode (output)
capacitance and the tune capacitor. Round conductors have a lower VHF
Q than flat conductors. To increase current handling ability, or to
reduce inductance, two paralleled round conductors, separated by a
wide air-gap, are better than a flat conductor of the same overall
width.
The simplest type of parasitic suppressor is a resistor. It
reduces Q by adding R. This technique is effective. However, it is
mostly limited to low power applications. The traditional
staggered-resonance parasitic suppressor provides two advantages over
a resistor suppressor--it can handle more current, and it causes the
VHF resonance to work against itself.
A staggered-resonance parasitic suppressor typically consists of a
coil inductor paralleled with a low-inductance resistor. The axis of
the coil inductor is parallel to the resistor. Here's how it works:
The magnetic field from current flowing in the resistor is at a right
angle to the direction of current flow. The magnetic field from the
inductor is parallel to the direction of current flow. Because the
two magnetic fields are 90 degrees apart, the inductances act
independently instead of mutually. The two independent-inductances
connect to one fixed-capacitance--i.e., the anode. Since the coil has
more inductance than the resistor, it creates a second VHF resonance
that is slightly lower in frequency than that produced by the
resistor. The conflicting resonances work against each other. This
technique broadbands the anode's VHF resonance, i.e., it reduces the
Q--very much like stagger-tuning IF transformers to widen a
receiver's band pass. Reducing VHF Q lowers the parallel equivalent
VHF load resistance on the anode. This reduces the VHF voltage
gain--and that reduces an amplifier's ability to oscillate at
VHF.
Choosing the optimum amount of inductance for a suppressor inductor
[Ls] is best determined experimentally by operating the amplifier on
10m. Since 10m is almost VHF, a device that suppresses VHF energy
should get hot from 10m RF. If Ls is too small, the suppressor
resistor [Rs] will not exhibit visible signs of heating during
operation on 10m. If Ls is too large, there will be too much voltage
drop on 10m and Rs will burn out.
Every straight conductor has inductance. The amount of inductance is
fairly proportional to length. Manufactured high-power
"non-inductive" resistors are long--and therefore somewhat inductive.
They are too inductive for use in VHF suppressors. However, it's easy
to make a suppressor resistor of sufficiently low inductance by
paralleling straight nichrome wires that are separated by
air-gaps.
Staggered-resonance suppressors can be built without a resistor by
paralleling two unequal-inductance nichrome wires. For example, a
silver plated strap in the anode circuit can be changed from a
potential source of grief into a Q-reducing asset by replacing it
with two, parallel, nichrome wire conductors. One of the conductors
is made about 25% longer than is necessary to span the distance. Its
length is shortened by winding a small 1 to 2 turn coil. The axis of
the coil is parallel to the shorter wire. This arrangement decouples
the magnetic field in the coil from the magnetic field in the
straight conductor.
For large amplifiers, 100% nichrome staggered-resonance suppressors
solve the problem of not being able to find a high-power resistor of
sufficiently low inductance for use in parasitic suppression service.
For very large HF amplifiers, all-nichrome staggered-resonance
suppressors should be made from flat nichrome conductors in order to
carry the large RF circulating current between the tune capacitor and
the anode capacitance.
In two-tube amplifiers, if the two suppressors are allowed to
magnetically couple to each other, a VHF parasitic oscillation may
occur. In a two-tube amplifier, the suppressor coils should be
positioned at a right angle. If the suppressor coils are parallel to
each other, the coils should be wound in opposite directions, and
separated as much as is practical.
Some amateur radio operators--and some electronic engineers--do
not believe that VHF oscillations can take place in an HF amplifier.
This is understandable because the most common and most destructive
type of VHF-parasitic-oscillation, the push-push variety, lasts only
a matter of microseconds. Push-pull VHF parasitic oscillation is
obviously only possible in multi-tube amplifiers. A steady
oscillation between anodes is the result. Push-pull parasitic
oscillation is characterized by extremely high anode dissipation,
moderate grid and anode currents with zero drive, and no arcing.
Push-pull parasitic oscillations can be stopped by switching the
amplifier to standby. This is not possible with a push-push parasitic
oscillation---wherein the event is probably over before the sound of
the parasitic arc reaches the operator's ears.
VHF parasitic oscillations are not cooperative. It may take a
particular sequence of anode-current transients to initiate a
parasitic oscillation. Even though there is no concrete scientific
evidence to prove it, the phrase "CQ contest" may have a propensity
to produce the key sequence of anode-current transients--especially
if the contest is one you've been waiting for, and the local radio
parts emporium has just closed for the weekend.
A major factor with parasitics is the VHF-gain of the particular
amplifier-tube or tubes that happen to be installed in the amplifier.
Even among new amplifier-tubes from the same production lot, there is
some variation in VHF gain. Tubes with below average VHF gain may
never have a parasitic-oscillation--no matter how poorly their
parasitic-suppressors perform. Thus, when below average gain tubes
happen to be installed in an amplifier, it's easy to assume that the
amplifier design is perfectly stable.
Since catching a parasitic oscillation in the act is virtually
impossible with ham-type test gear, a different analytical approach
must be used. It's a reasonable assumption that a resonant circuit
which supports parasitic oscillation can be found and evaluated with
a dipmeter.
To determine the parasitic frequency and evaluate the parasitic
suppressor, unplug the HF amplifier from the electric-mains and
measure the anode-resonance with a dipmeter. The best place to do
this is on either side of the HV blocking capacitor. The resonant
frequency typically varies inversely with the amplifier's power
capability. 700W single-tube amplifiers typically resonate from
100MHz to 150MHz. 1500W amplifiers typically resonate from 80MHz to
140MHz. 100kW amplifiers typically resonate from 35MHz to 45MHz. You
should be able to tune the resonance a few MHz by adjusting the tune
capacitor. The VHF dip in some amplifiers' anode-circuits is so sharp
that it will "suck-out" the oscillator in the dipmeter. If that is
the case, the dipmeter must be backed away (decoupled) from the
conductor to accurately observe the dip. A broad, smooth dip is good.
A sharp dip indicates that the anode-circuit has a high VHF-Q. This
is not good news unless you happen to need a VHF oscillator.
If the suppressor design is changed to (hopefully) lower the VHF Q,
check the dip again. The frequency will usually not change
appreciably but the dip should now be smoother, more broad, and it
should be necessary to couple the dipmeter coil closer to the
anode-circuit to achieve the same degree of dip.
If you make an experimental change to a parasitic suppressor, and you
want to evaluate the change more precisely, use a plastic ruler to
measure the distance from the anode circuit to the tip of the
dipmeter coil that will result in a 20% dip on the dipmeter. If the
coupling distance required for the 20% dip decreases after a change
to a suppressor, a lower VHF-Q is indicated and the change was
obviously an improvement. If the dip distance increases, the VHF-Q
went up and the change in the suppressor was a step backward.
Amplifiers that exhibit a tuning vagary in the area of resonance on
one or two bands are probably in need of better parasitic
suppression. A stable amplifier usually exhibits smooth, symmetrical
tuning.
Additional information on parasitic oscillation was published in the
September and October 1990 issues of QST.
END OF PART 3
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